# The Automatic Gain Control: topology, behavior and use (I)

One of the most common topologies in electronic design is the Automatic Gain Control (AGC). In this post we will study what is its operating modes, its basic topology and its most common use. We will also make to its simulation in MatLab, using SIMULINK, understanding better its behavior.

LINEAR AMPLIFIERS

One of the most common blocks in an electronic system is the linear amplifier. This is a device where the output signal is directly proportional to the input. As the output level is greater than the input, the block increase the signal level, then it is an amplification. If the output level was lower than the input level, then we would speak of a level reduction or attenuation.

The linear amplifiers usually have a fixed gain, which is the proportionality constant between the input and the output signals, and a variable gain, where this ratio can be controlled an external control voltage vc.

$v_{out}(t)=g v_{in}(t)$
$v_{out}(t)=g(v_c(t)) v_{in}(t)$

This voltage is a variable that also depends on time, although under conditions of free control, done by the user, once the control value is chosen, this variable becomes stationary with time and the amplifier becomes fixed gain.

However, the input signals may have oscillations due to the propagation channel, and increase or decrease in value as a function of time. If the amplifier has a fixed gain, the output will follow lineary the input variations.

In general, conventional amplifiers usually have a fixed gain with an external regulation that can be controlled by the user. However, within communication systems there may be cases in which it is always necessary to ensure that the output takes a fixed value. And for this it is needed to use an Automatic Gain Control (AGC).

THE AUTOMATIC GAIN CONTROL

The AGC is a feedback system, which uses the output variable, taking a sample, and processes it properly generating a control voltage vc(t) that varies the output level, keeping it fixed against the input variations.

The typical AGC block diagram can be seen in the following figure

Fig. 1 – Block diagram for an AGC

It consists of a variable voltage amplifier, which is expressed by the formula seen in the previous section, an envelope detector, because the amplitude of the signal vout contains the information of the variation of the input signal, since vout is proportional a vin, a comparator, which compares the detected signal with a reference signal vref, which is the one that will govern the appropriate output level in vout and an integrating filter, which provides the control variable.

By varying vin at time t0, the VGA is in a steady state, behaving like a linear gain amplifier. This causes a variation in the output signal vout that follows the input vin. This variation is detected by the envelope detector, causing a change in the comparator output, which, when integrated, modifies the value of vc, adapting it so that vout is corrected and starts to match the value before the change.

It is a dynamic process: the vin and vout signals vary temporarily but keeping a stationary envelope level. For example, a pure sine wave has a constant envelope varying in [-1, 1] intervale

Fig. 2 – Sinewave

When a change in the envelope is detected at a certain time, the peak value of the amplitude changes and is detected by the detector, which initiates a temporary feedback process that does not affect the waveform, but does affect its amplitude.

Fig.3 – Change in the sinewave amplitude

CONTROL MECHANISMS IN AN AGC

Returning to the system in Fig. 1, where the VGA has a gain represented by the expression

$g(v_c(t))=g_o e^{-\alpha v_c(t)}$

In this expression the temporal domain is removed, because at this moment we are not interested in the temporal variation of vc, since if there is no variation in vin, vc remains stationary.

The input signal is the next

$v_{in}(t)=a \sin({\omega}t+{\theta})$

and the output signal is

$v_{out}(t)=g_o a e^{-\alpha \cdot v_c(t)} \sin({\omega}t+{\theta})$

This signal will pass through the envelope detector, the output of which is a signal that is proportional to the amplitude of the input signal, where k is the proportionality constant. Therefore, the output signal of the envelope detector will be

$v_e=k g_o a e^{-\alpha v_c(t)}$

This signal is passed through a logarithmic amplifier, since the dependence of vE on vc is exponential. Since the base is natural, we choose the natural logarithm as the logarithmic amplifier, and we can get an output voltage v2 whose expression is

$v_2=-{\alpha} v_c+\log(k g_o a)$

In this expression we can verify that k and g0 are constant values, and that a and vc are the ones that can vary with respect to time. If we now include the temporal variation of a, we will have that the expression is

$v_2=-{\alpha} v_c(t)+\log(k g_o a(t))$

Therefore a variation of a is compensated by a variation of vc so that v2 returns to the value before the change in a.

Making the comparison between the voltage v2(t) and vR, which is a fixed value and that will fix the output level on the amplifier, we have a signal v1 that has the following expression

$v_1 = - {\ alpha} v_c (t) + \ log (k g_o a (t) e ^ {- v_R})$

This signal is passed through a low-pass filter that integrates it, getting vC(t). If the filter has a transfer function h(t), what we do is a convolution of the signal v1 with h(t)

$v_c(t)=h(t)*v_1(t)$

And then

$v_1(t)+{\alpha} h(t)*v_1(t)=\log(k g_o a(t) e^{-v_R})$

In the temporal domain, convolution is a dynamic integral equation, so if we use the Laplace domain, we will transform that convolutional response to a response in the domain of the complex variable s, which is linear. Using this domain, the equation above is now

$V_1 (s) + {\alpha} H (s) V_1 (s) = \mathcal {L} [\log (k g_o a (t) e ^ {- v_R})]$

which is the Laplace transform. Studying the value of V1(s) if the output has a value an amplitude b

$v_ {out} (t) = b \sin ({\omega} t + {\theta})$

removing the dependency with k and with g0. Thus, making the same steps as in the previous case, we will have to

$v_1 (t) = \log (b (t) e ^ {- v_R})$

$V_1 (s) = \mathcal {L} [\log (b (t) e ^ {- v_R})]$

$(1 + {\alpha} H (s)) \mathcal {L} [\log (b (t) e ^ {- v_R})] = \mathcal {L} [\log (k g_o a (t ) e ^ {- v_R})]$

$\dfrac {\mathcal {L} [\log (b (t) e ^ {- v_R})]} {\mathcal {L} [\log (k g_o a (t) e ^ {- v_R}) ]} = \dfrac {1} {1 + {\alpha} H (s)}$

The first term is the quotient of two functions, one of them depends on the output amplitude and the other depends on the input amplitude. If we choose k · g0 = 1, we will get

$\dfrac {\mathcal{L}[\log(b(t) e^{-v_R})]}{\mathcal{L}[\log(a(t) e^{-v_R})]}=\dfrac {\mathcal{L}[\log(b(t))]}{\mathcal{L}[\log(a(t)]}=\dfrac {1}{1+{\alpha} H(s)}$

And being y(t) and x(t) voltage values, we can apply the dB definition, which is

$b_{dB}(t)=20 \log_{10}(b(t))$

$a_{dB}(t)=20 \log_{10}(a(t))$

and then

$\dfrac {\mathcal{L}[\log(b(t) e^{-v_R})]}{\mathcal{L}[\log(a(t) e^{-v_R})]}=\dfrac {\mathcal{L}[b_{dB}(t)]}{\mathcal{L}[a_{dB}(t)]}=\dfrac {B_{dB}(s)}{A_{dB}(s)}$

removing the temporary domain and turning the system into a totally linear system. Then we will have to

$\dfrac {B_{dB}}{A_{dB}}=\dfrac {1}{1+{\alpha} H(s)}$

being the transfer function in dB of the variation between the output and input amplitudes.

If the filter used is an integrating filter with a pole at the origin (low-pass filter), like this

$H(s)= \dfrac {C}{s}$

the expression will be

$\dfrac {B_{dB}}{A_{dB}}=\dfrac {1}{1+{\alpha} C}$

Now suppose that the input envelope AdB changes 1 dB, increasing or decreasing. The new envelope is A’dB(s), and the output envelope, B’dB(s). Then:

${A'}_{dB}(s)=A_{dB}(s) \pm 1$

And having

$\dfrac {B_{dB}}{A_{dB}}=\dfrac {{B'}_{dB}}{{A'}_{dB}}=\dfrac {1}{1+{\alpha} C}$

since feedback must always respond in the same way. Substituting the expression for the input variation in the previous expression we have

$\dfrac {B_{dB}}{A_{dB}}=\dfrac {{B'}_{dB}}{A_{dB}(s) \pm 1}=\dfrac {1}{1+{\alpha} C}$

Se we can calculate B’dB(s) multiplying by the transfer function

${B'}_{dB}(s)=\dfrac {s}{s+{\alpha} \cdot C} \cdot A_{dB}(s) \pm \dfrac {s}{s+{\alpha} C}$

And knowing that the first term is BdB(s), the expression will be the next

${B'}_{dB}(s)-B_{dB}(s)=\pm \dfrac {s}{s+{\alpha} C}=\pm 1 \mp \dfrac {{\alpha} C}{s+{\alpha} C}$

The above equation links the new envelope B’dB(s) with the former BdB(s). And being a transient response, applying the inverse transformation it is got

${B'}_{dB}(t)-B_{dB}(t)=\pm {\delta}(t) \mp {{\alpha} C e^{-{\alpha} C t}}$

Lt’s study this result: When 1 dB (instant t=0) is increased, the expresion is b’dB(t)–bdB(t)=+δ(t)=+1, because at t=0 the filter h(t) has not worked yet. Therefore, at this time the difference between the new and the initial envelope is 1 dB. When t is increasing, there is an decreasing exponential response due to the second term of the previous expression, ans when the time is increasing more, the difference between b’dB(t) and bdB(t) is decreasing (inicially b’dB(t)>bdB(t)) until both are equal.

Conversely, decreasing the input envelope 1dB, then b’dB(t)–bdB(t)=-δ(t)=-1, and the final envelope decreases this value for the same reason, and the operation is the inverse of the previous case.

From this it follows that when the input envelope rises or falls 1 dB, the output envelope, at the initial moment, tends to rise or fall following the variation of the input envelope, but when time passes, the output envelope stabilizes until it reaches the initial value ydB(t).

The AGC time response is α·C/e es τ=1/α·C, which is a constant time. When this constant is high, the AGC changes slowly, but being low, the AGC changes fastly. A compromise with the AGC response time is required in signals that also contain nominal variations for their content, such as analogue audio or video signals, so as not to confuse a level variation with a variation of that content.

CONCLUSION

In this entry we have been able to verify what is the theoretical behavior of the AGC block diagram, studying its response in the Laplace domain and in the temporal domain. We have reached a transfer relationship that allows us to relate the variations of the output signal to the input signal and how we can calculate the AGC response time, which we will have to include through the integrating filter and the study of the variation constant of the amplifier gain.

In the next post we will study this system using SIMULINK.

REFERENCES

1. Benjamin C. Kuo; “Automatic Control Systems”; 2nd ed.; Englewood Cliffs, NJ; Prentice Hall; 1975
2. Pere Matí i Puig; “Subsistemas de radiocomunicaciones analógicos”;Universitat Oberta de Catalunya;2010

# Basis of Microwave Heating

Microwave oven has become very popular in recent years, and has become an essential appliance in any kitchen. However, microwave heating seems an esoteric, almost magical, issue for many people who have the oven at their home. In this post we are going to explain the basis of microwave heating, not only for the food heating, but also for industrial heating and HDW (hot domestic water).

In 1946, a British researcher from the Raytheon Corporation, Mr. Percy Spencer, working on RADAR applications, discovered that a candy bar in his pocket was melted. He was testing a magnetron and began experimenting, confining the EM field inside a metal cavity. He tested first with corn and then with a chicken egg. This latter one exploded.

He verified that a high intensity EM field affected food due to the presence of water inside. Water is a bad propagator of radio waves, because it has a high dielectric constant and losses. Being a polar molecule, when a variable EM field is applied, the dipole tends to be oriented in the direction of the field, and that makes the water molecule is agitated, increasing its temperature. The popular belief is that this only happens at 2,4 GHz, but it actually happens throughout the microwave band. This frequency is used by the ovens because it is a frequency within a free emission band known as ISM (short for Industrial, Scientific and Medical). However, there are heating processes at 915MHz and another frequencies..

First, the water, like almost all dielectrics, has under normal conditions a complex dielectric constant ε=ε’−jε”. When this complex dielectric constant is introduced into the Maxwell equations, the complex term means a dielectric conductivity, by the next expression

$\sigma = \omega \epsilon" \epsilon_0$

This conductivity is not produced by the mobility of electrons, but by the mobility of the polar molecules of water. Therefore, it is higher as the frequency is increased.

On the other hand, the presence of this conductivity limits the microwaves penetration in the water, attenuating the EM intensity with distance. It is related to the depth of penetration, expressed by

$\delta_p=\dfrac {\lambda \sqrt{\epsilon'}}{2 \pi \epsilon"}$

and therefore at higher frequency, lower penetration depth. If the intensity of the electric field is |E|, and, by the Ohm’s law, the volumetric power is given by

$Q=\omega \epsilon" \epsilon_0 |E|^2$

This volumetric power will affect a specific region of the water, causing heating.

On the other hand, there is a heat transfer effect due to thermal conductivity, such that the surface heat flux is

$\dfrac {dQ_s}{dt}=-k \displaystyle \int_s {\vec \nabla T d \vec S}$

Applying the divergence theorem, the variation of heat per unit volume will be

$\dfrac {dQ_V}{dt}=-k \nabla^2 T$

This flow distributes the temperature inside the volumetric element, lossing energy, and therefore its sign is negative.

WATER HEATING

Under macroscopic conditions, the energy per unit volume that must be applied to water increasing its temperature is given by

$E_v=\rho_m c_e \Delta T$

with ρM the water density and ce its specific heat, being ΔT the increasing of the temperature. Speaking in terms of power, we will have to

$Q=\rho_m c_e \dfrac{dT}{dt}$

where it must be calculate the global time variation of the temperature, and being a fluid that can be in movement, it must be applied the material derivative, an operator that includes the time variation and the convection. Applying this operator we may get

$\dfrac{dT}{dt}=\dfrac{\partial T}{\partial t}+\vec v \vec \nabla T$

and the volumetric power is given by

$Q=\rho_m c_e \left(\dfrac{\partial T}{\partial t}+\vec v \vec \nabla T \right)-k\nabla^2 T$

which is the expression that governs the water heating when a volumetric density of EM power Q is applied.

On the other hand, fluid movement is governed by the Navier-Stokes equations, through

$\rho_M \dfrac {\partial \vec v}{\partial t}=-\vec \nabla P+\mu \nabla^2 \vec v + \rho_M \vec g$

Where P is the volumetric pressure, μ the fluid viscosity and g the gravitational field.

HDW SYSTEMS USING MICROWAVES

In the case of a hot domestic water system, there would be two possibilities of heating:

1. Through a closed circuit system moving a water flow, due to its very low viscosity (10-3 Pa·s).
2. Using a vessel with rest water and accumulating the heat to transmit it to another areas.

In the first case, the volumetric power necessary to heat a closed circuit system must solve both with the thermal variation and the Navier-Stokes equations, and its efficiency is greater than in the second one, where the expression of the thermal increase is given by

$Q+k\nabla^2 T=\rho_m c_e \dfrac{\partial T}{\partial t}$

This equations can be solved using the FEM method, as we saw in the post about the simulation.

In any case, although both methods are possible, the first method will always be cheaper than the second, since the second can only be applied to raise the temperature of another fluid in motion and will need more energy due to the losses due to that transfering of heat..

IS IT POSSIBLE TO HEAT OTHER MATERIALS USING MICROWAVES?

Normally, any material that has losses by dielectric constant can be capable of being heated using microwaves, if these losses do not raise the electrical conductivity to values that cancel the electric field (in a perfect conductor, the electric field is zero). If we write the expression obtained in terms of electric field we get

$\omega \epsilon" \epsilon_0 |E|^2+k\nabla^2 T=\rho_m c_e \left(\dfrac{\partial T}{\partial t}+\vec v \vec \nabla T \right)$

and therefore, we can obtain a relationship between ε” and the increase of temperature at a given electric field |E|.

INFLUENCE ON THE HUMANS

The human body is another dielectric which contains mostly by water. Therefore, the effect of the EM radiation on our body should cause heating. Let’s study what would be the field that would increase our temperature above 50o C in one minute, reducing the expressions to

$\omega \epsilon" \epsilon_0 |E|^2=\rho_m c_e \dfrac{\Delta T}{\Delta t}$

Taking ε”=4,5 (water at 2,4 GHz), knowing that the average human density is 1100 kg/m3 and its specific heat, 14,23 kJ/kg o C, it is got the next

$|E|=\sqrt {\dfrac {1100 \cdot 14230 \cdot \left(\dfrac{50-33}{60} \right)}{2 \pi \cdot 2,4 \cdot 10^9 \cdot 4,5 \cdot 8,85 \cdot 10^{-12}}}=3,1 kV/m$

and a WIFI router emits with less than 2 V/m field strength at 1 m. of distance. Therefore, a WIFI router will not cause heating in our body or even if we are close by it..

And a mobile phone? These devices are already powerful … Well, at its emission peak either, since at most it will emit with 12 V/m, and we need 3100 V/m, about 260 times more. So the mobile does not warm our ear either. And keeping in mind the depth of penetration, as much the EM radiation gets to penetrate about 2 cm, attenuating the field strength in half and power to the fourth part, due to the dielectric conductivity of our body. That without keeping in mind that each of our tissues has a different attenuation capacity depending on its composition and structure.

CONCLUSION

This post tries to explain the microwave heating phenomenon based on the ones that produce this heating, and its possible industrial applications, apart from those already known as the popular oven that almost every kitchen already has as part of its home appliance furniture. One of the most immediate applications is in the HDW, although applications have also been achieved in other industrial areas. And although the microwaves produce that heating, the necessary field strengths are very far from the radiation we receive from mobile communications.

REFERENCES

1. Menéndez, J.A., Moreno, A.H. “Aplicaciones industriales del calentamiento con energía microondas”. Latacunga, Ecuador: Editorial Universidad Técnica de Cotopaxi, 2017, Primera Edición, pp 315. ISBN: 978-9978395-34-9
2. D. Salvi, Dorin Boldor, J. Ortego, G. M. Aita & C. M. Sabliov “Numerical Modeling of Continuous Flow Microwave Heating: A Critical Comparison of COMSOL and ANSYS”, Journal of Microwave Power and Electromagnetic Energy, 2016, 44:4, 187-197, DOI: 10.1080/08327823.2010.11689787

# Simulation on Physical Systems

I take a long time writing many post about the simulation. Main reason is because I have learned for many years the value of using computers for physical system analysis. Without these tools, I would never be able to get reliable results, because of the amount of calculations I would have to do. Modern simulators, able to solve complex calculations using the computers capacity, allow us to get a more realistic behavior for a complex system, knowing its structures. Physics and Engineering work every day with simulations to get better predictions and take decisions. In this post, I am going to show what are the most important parts we should be kept in mind about the simulation.

In 1982, physicist Richard Feynman published an article where he talked about the analysis of physical systems using computers (1). In those years, computer technology had progressed to a high level that it was possible to achieve a greater calculation capacity. New programming languages worked with complex formulas, such as FORTRAN, and allowed the calculations on systems by complex integro-differential equations, which resolution usually needed numerical methods. So, in those first years, physicists began to do simulations with programs able to solve the constitutive system equations, although not always with simple descriptions.

A great step forward in electronics was the SPICE program, at the beginning of 70s (2). This program, FORTRAN-based, was able to compute non-linear electronic circuits, removing the radiation effects, and solve the time-domain integral-differential equations. Over the years, the Berkeley’s SPICE became the first reference on simulation programs and its success being such that almost all the simulation programs developed along last years have its base on the Nagel and Pederson algorithms, developed in 70s.

From 80s, and searching to solve three-dimensional problems, the method of moments (MoM) was developed. It was come to solve systems raised as integral equations in the boundaries (3), being very popular. It was used in Fluid Mechanics, Acoustic Waves and Electromagnetism. Today, this one is still used to solve two-dimensional electromagnetic structures.

But the algorithms have got a huge progress, with the emergence of new finite element methods (FEM, frequency-domain) and time-domain finite differences (FDTD, time-domain) in 90s, based on the resolution of systems formulated by differential equations, important benchmarks on the generation of new algorithms able to solve complex systems (4). And with these new advances, the simulation contribution in Physics came to take spectacular dimensions.

WHAT IS THE VALUE OF AN ACCURATE MODEL?

When we are studying any physical phenomenon, we usually invoke a model. Whether an isolated phenomenon or within an environment, whether in Acoustic Waves, Electromagnetism or Quantum Mechanics, having a well-characterized model is essential to get its behavior, in terms of its variables. Using an accurate model increases our certainty on the results.

However, modeling is complex. It is needed to know what are the relationships between variables and from here, determine a formulation system that defines the behavior within a computer.

A model example is a piezoelectric material. In Electronics, piezoelectric materials are commonly used as resonators and it is usually to see these electronic devices (quartz or any other resonant material based on this property).

A piezoelectric model, very successful in the 40s, was developed by Mason (5). Thanks to the similarity between the Electromagnetic and Acoustic waves, he got to join both properties using transmission lines, based in the telegraphist’s equations, writing the constitutive equations. In this way, he developed a piezoelectric model which is still used today. This model can be seen in Fig. 1 and it has already been studied in previous posts.

Fig.1 – Modelo de piezoeléctrico de Mason

This model practically solved the small signal analysis in frequency domain, getting an impedance resonance trace as it is shown in Fig. 2

Fig.2 – Resultados del análisis del modelo de Mason

However, the models need to expand their predictive capacity.

The Mason model describes the piezoelectric behavior rightly when we are working in a linear mode. But it has faults when we need to know the large signal behavior. So new advances in the piezoelectric material studies included the non-linear relationships in its constitutive equations (6).

Fig. 3 – Modelo tridimensional de una inducción

In three-dimensional models, we must know well what are the characteristics that define the materials to have an optimal results. In the induction shown in Fig. 3, CoFeHfO is being used as a magnetic material. It has a frequency-dependent complex magnetic permeability that must be defined in the libraries.

The results will be better as the model is defined better, and this is the fundamental Physicist task: getting a reliable model from the studies on the phenomena and the materials.

The way to extract a model is usually done by direct measurement or through the derived magnitudes, using equations systems. With a right model definition, the simulation results will be more reliable.

ANALYSIS USING SIMULATION

Once the model is rightly defined, we can perform an analysis by simulation. In this case, we will study the H-field inside the inductor, at 200 MHz, using the FEM analysis, and we are going to draw this one, being shown in Fig. 4.

Fig. 4 – Excitación magnética en el interior del inductor

The result is drawn in a vector mode, since we have chosen that representation to see the H-field direction inside the inductor. We can verify, first, that the maximum H-field is inside the inductor, to the positive section on Y axis in the upper area, while in the lower part the orientation the inverse. The maximum H-field level obtained is 2330 A/m with 1 W excitation between the inductor electrodes.

The behavior is precisely that of an induction whose value can also be estimated by calculating its impedance and drawiing it on Smith’s chart, Fig. 5.

Fig. 5 – Impedancia del inductor sobre carta de Smith

The Smith’s chart trace clearly shows an inductive impedance, which value decreases when the frequency increases, because of losses of the CoFeHfO magnetic material. Besides, these losses contribute to the resistance increasing with frequency. There will be a maximum Q in the useful band

Fig. 6 – Factor de calidad del inductor

Having a induction with losses a quality factor Q, we can draw it as a function of the frequency in Fig. 6.

Therefore, with the FEM simulation we have been able to analyze the physical parameters on a modeled structure that would have cost us much more time and effort to get by means of complex calculations and equations. This shows, as Feynman pointed out in that 1982 conference, the simulation powerful when there are accurate models and proper software to perform these analyzes.

However, the simulation has not always had the chance to get the best results. Precisely is the previous step, the importance of having an accurate model, which faithfully defines the physical behavior of any structure, which will ensure the reliability of the results.

EXPERIMENTAL RESULTS

The best way to check if the simulation is valid is to resort getting experimental results. Fortunately, the simulation performed on the previous inductor is got from (7), and, in this reference, the authors show experimental results that validate the results of the inductor model. In Fig. 7 and 8 we can see the inductance and resistance values, and adding the quality factor, can be compared with the experimental results of the authors.

Fig. 7 – Valor de la inductancia en función de la frecuencia

Fig. 8 – Valor de la resistencia efectiva en función de la frecuencia

The results obtained by the authors, using HFSS for the simulation of the inductor, can be seen in Fig. 9. The authors have done the simulation on the structure with and without core, and show the simulation against the experimental result . Seeing the graphs, it can be concluded that the results got in the simulation have a high level of concordance with those obtained through the experimental measurements.

This shows us that the simulation is effective when the model is reliable, and that a model is accurate when the results obtained through the simulation converge with the experimental results. In this way, we have a powerful analysis tool that will allow us to know in advance the behavior of a structure and make decisions before moving on to the prototyping process.

In any case, convergence is also important in a simulation. The FEM simulation needs that the mesh is so accurate as getting a good convergence. A low convergence level gives results far from the optimum, and very complex structures require a lot of processing speed, a high RAM use and, sometimes, must even perform a simulation on several processors. To more complex structures, the simulation time increases considerably, and that is one of its main disadvantages.

Although the FEM simulators allow the optimization of the values ​​and even today the integration with other simulators, they are still simulators that require, due to the complexity of the calculations to be carried out, powerful computers that allow to make those calculations with reliability.

CONCLUSIONS

Once again, we agree with Feynman when, in that 1982 seminar, he chose precisely a topic which seemed to have no interest for the audience. Since that publication, Feynman’s article has become a classic of Physics publications. The experience that I have got over the years with several simulators, shows me that the way opened by them will have a considerable advance when quantum computers are a reality and their processing speed raises, allowing that these tools get reliable results in a short space of time.

The simulation in the physical systems has been an important progress to get results without needing to realize previous prototypes and supposes an important saving in the research and development costs.

REFERENCES

1. Feynman, R; “Simulating Physics with Computers”; International Journal of Theoretical Physics, 1982, Vols. 21, Issue 6-7, pp. 467-488, DOI: 10.1007/BF02650179.
2. Nagel, Laurence W. and Pederson, D.O. “SPICE (Simulation Program with Integrated Circuit Emphasis)”, EECS Department, University of California, Berkeley, 1973, UCB/ERL M382.
3. Gibson, Walton C., “The Method of Moments in Electromagnetics”, Segunda Edición, CRC Press, 2014, ISBN: 978-1-4822-3579-1.
4. Reddy, J.N, “An Introduction to the Finite Element Method”, Segunda Edición,  McGraw-Hill, 1993, ISBN: 0-07-051355-4.
5. Mason, Warren P., “Electromechanical Transducers and Wave Filters”, Segunda Edición, Van Nostrand Reinhold Inc., 1942, ISBN: 978-0-4420-5164-8.
6. Dong, S. Shim and Feld, David A., “A General Nonlinear Mason Model of Arbitrary Nonlinearities in a Piezoelectric Film”, IEEE International Ultrasonics Symposium Proceedings, 2010, pp. 295-300.
7. Li, LiangLiang, et al. 4, “Small-Resistance and High-Quality-Factor Magnetic Integrated Inductors on PCB”, IEEE Transactions on Advanced Packaging, Vol. 32, pp. 780-787, November 2009, DOI: 10.1109/TADVP.2009.2019845.

# Studying slotline transmission lines

PCB transmission lines are an optimal and low cost solution to make guided propagation at very high frequencies. The most popular lines are microstrip and coplanar waveguide. These transmission lines are easily realizable in a printed circuit board and whose impedance can be calculated from their dimensions. In these lines, TEM modes (transverse electromagnetic) are propagated, in which there is no component in the direction of propagation. However, there are other very popular lines that can also be used at high frequencies and are known as slotlines. In this post, we are going to study the electrical behavior of slotlines and some microwave circuits that can be done with them.

At high frequencies, lines usually behave like distributed transmission lines. Therefore, it is necessary to know its impedance so that there are no losses during propagation.

The microstrip and coplanar waveguides are very popular, since they are easily implemented on a printed circuit board, they are cheap and can be easily calculated. In both lines, the propagation mode is TEM, there are no field components in the direction of propagation, and their characteristic impedance Zc and wavelength λg depend on the line dimensions and the dielectric substrate which supports them.

There is another type of line, which is usually used at very high frequencies: the slotline. This line is one slot on the copper plane through which a transverse electric mode is propagated (specifically the TE01 mode, as shown in the following figure).

Fig. 1 –  TE01 mode on a slotline

The field is confined near the slot so that the propagation has the minimum possible losses, and as the microstrip lines, there is a discontinuity due to the dielectric substrate and air. It is used as a transmission line with substrates with a high dielectric constant (around εr≥9.2), in order to confine the fields as close as possible to the slot, although they can be used as couplings on substrates with lower dielectric constants. In this way, flat antennas can be fed with the slotlines.

In this post, we will pay attention to its use as transmission lines (with high dielectric constants), and the microwave circuits that we can make with them, studying the transitions between both technologies (slotline to microstrip).

ANALYZING THE SLOTLINE TRANSMISSION LINE

Being a transmission line and like the other lines, the slotline has a characteristic impedance Zc and a wavelength λs. But besides, using the TE01 propagation mode, the electric field component which is propagated, in cylindrical coordinates, is Eφ, as it is shown in the next figure

Fig. 2 – Eφ component

This component is calculated from the magnetic components Hr and Hz, considering the Z-axis the propagation direction, which is perpendicular to the electric field. From here, we get an expression for the propagation constant kc which is

$E_{\varphi}=\dfrac {j{\omega}{\mu_0}}{k_c^2}\dfrac {\partial H_z}{\partial r}=-{\eta} \dfrac {\lambda_s}{\lambda_0}H_r$

$k_c=\dfrac {2{\pi}}{\lambda_0} \sqrt {1- \left( \dfrac {\lambda_0}{\lambda_s} \right)^2}$

where λ0 is the wavelength of the propagated field. The first thing is deduced from the expression of kc is that we will find a cuttoff wavelength λs, from which the field propagates as mode TE01, since λ0≤λs so that kc is real and there is propagation. This means that there will be a cuttoff thickness for the substrate which will depend on the dielectric constant εr. The expression for that cuttoff thickness, where there is no propagation at TE01 mode, is

${\left( \dfrac {h}{\lambda_0} \right)}_c=\dfrac {1}{4\sqrt{{\epsilon_r}-1}}$

With these expressions, Gupta (see [1], page 283) got the expressions for the line impedance Zc and the line wavelength λs, which will allow us to typify the transmission line, making microwave circuits with slotlines.

ANALYZING A SLOTLINE

As the microstrip and coplanar waveguides, slotline can be analyzed using a FEM electromagnetic simulator. We are going to study one transmission line on an RT/Duroid 6010 substrate, which dielectric constant is εr=10,8, with 0,5mm thickness. The slot width is 5mil. According to the impedance calculations, Zc is 68,4Ω and λs, 14,6mm, at 10GHz. In a 3D view, the slotline is the next

Fig. 5 – Slotline 3D view

The next graph shows the S parameters at 50Ω impedance of generator and load.

Fig. 6 – Slotline S parameters

On the Smith chart

Fig. 7 – Slotline impedance on Smith Chart

where the impedance is 36,8-j·24,4Ω at 10GHz.

It is possible to show the propagated surface current on the line in 3D view

Fig. 8 – Slot surface current, in A/m

where it can be seen that the surface current is confined as near as possible the slot. From this current, the H-field can be derived and therefore the E-field which only has a transversal component. It can be also seen two maxima on the current magnitude, which shows that the slot distance is λs.

The FEM simulation allows us to analyze the slotline lines and build microwave circuits, knowing the characterization shown in [1].

SLOTLINE-TO-MICROSTRIP TRANSITIONS

Like the slotline is one slot made on a copper plane, transitions can be made from slotline to microstrip. One typical transition is the next

Fig. 9 – Slotline-to-microstrip transitions

Microstrip lines finish in a λm/4 open circuit stub, so the current is minimal at the open circuit and maximum at the transition location. In the same way, the slotline finishes in a λs/4 short circuit stub, with the minimum surface current at the transition location. Then, the equivalent circuit for each transition is

Fig. 10 -Equivalent circuit for a slotline-to-microstrip transition

Using the FEM simulator it is possible to study how a transition behaves. The next graph shows its S parameters. The transition has been made on RT/Duroid 6010, with 70mil thickness and 25mil slot widths. The microstrip width is 50mil and the working band is 0,7÷2,7GHz.

Fig. 11 – Transition S parameters

and showing the surface current on the transition, it ts the next

Fig. 12 – Current on the transition.

where it can be seen the coupling of the current and its distribution on the slotline.

ANOTHER MICROWAVE CIRCUITS BASED ON SLOTLINES

The slotline is a versatile line. Combined with microstrip (the microstrip ground plane can include slots), it allows us to make a series of interesting circuits, such as those shown in fig. 13

Fig. 13 – Microwave circuits with slotline and microstrip.

The 13 (a) circuit shows a balum with slotline and microstrip technology, where the microstrip is shorted to ground in the transition. The balanced part is the slotline section, since both ground planes are working like differential ports, while the unbalanced part is the microstrip, referring to the ground plane where the slots are placed. With this circuit it is possible to build frequency mixers or balanced mixers. Another interesting circuit is shown in 13 (b), a “rat-race” where the microstrip circuit is not closed, but is coupled through a slot to get the coupling. In 13 (c), a “branchline” coupler is shown, using a slotline and, finally, in 13 (d), a Ronde coupler is shown. This last circuit is ideal to equalize the odd/even mode phase velocities.

CONCLUSIONS

In this post, we have analyzed the slotline used like a microwave transmission line, compared with another technologie. Besides we have made a small behavior analysis using an FEM simulator, checking the possibilities of the line analysis (S parameters and surface current analysis) and we have shown some circuits that can be made with this technology, verifying the versatility of this transmission line.

REFERENCES

1. Gupta, K.C., et al. “Microstrip Lines and Slotlines”. 2nd. s.l. : Artech House, Inc, 1996. ISBN 0-89006-766-X.

# Simulating transitions with waveguides

Waveguides are transmission lines widely used in very high frequency applications as guided propagation devices. Their main advantages are the reduction of losses in the propagation, due to the use of a single conductor and air, instead of using dielectrics as in the coaxial cable, a greater capacity to use high power and a simple building. Their main drawbacks are usually that they are bulky devices, that they cannot operate below their cutoff frequency and that the guide transitions to other technologies (such as coaxial or microstrip) have often losses. However, finite element method (FEM) simulation allows us to study and optimize the transitions that can be built with these devices, getting very good results. In this post we will study the waveguides using an FEM simulator such as HFSS, which is able to analyze tridimensional electromagnetic fields (3D simulation).

Waveguides are very popular in very high frequency circuits, due to the ease of their building and their low losses. The propagated fields, unlike the coaxial guides, are electric or magnetic transverse (TE or TM fields), so they have a magnetic field component (TE) or electric field (TM) in the propagation direction. These fields are the solutions for the Helmholtz equation under certain boundary conditions

• For the TE modes, Ez(x,y)=0

$\left( \dfrac {{\partial}^2}{\partial x^2} +\dfrac {{\partial}^2}{\partial y^2} +k_c^2\right)H_z(x,y)=0$

• For the TM modes, Hz(x,y)=0

$\left( \dfrac {{\partial}^2}{\partial x^2} +\dfrac {{\partial}^2}{\partial y^2} +k_c^2\right)E_z(x,y)=0$

and solving these differential equations by separation of variables, and applying the boundary conditions of a rectangular enclosure, where all the walls are electrical walls (conductors, in which the tangential component of the electric field is canceled)

Fig. 2 – Boundary conditions on a rectangular waveguide

we can obtain a set of solutions for the electromagnetic field inside the guide, starting from the solution obtained for the expressions shown in fig. 1.

Fig. 3 – Table of electromagnetic fields and parameters in rectangular waveguides

Therefore, electromagnetic fields are propagated like propagation modes, called TEmn, for the transverse electric (Ez=0), or TMmn, for the transverse magnetic (Hz=0). From the propagation constant Kc is got an expression for the cutoff frequencyfc, which is the lowest frequency for propagating fields inside the waveguide, which expression is

$f_c=\dfrac {c}{2} \sqrt {\left( \dfrac {m}{a} \right) ^2+\left( \dfrac {n}{b} \right) ^2}$

The lowest mode is when m=0, since although the function has extremes for m,n=0, the modes TE00 or TM00 do not exist. And like a>b, the lowest cutoff frequency of the waveguide is for the mode TE10. That is the mode we are going to analyze using a 3D FEM simulation.

SIMULATION OF A RECTANGULAR WAVEGUIDE BY THE FINITE ELEMENTS METHOD

In a 3D simulator it is very easy to model a rectangular waveguide, since it is enough to draw a rectangular prism with the appropriate dimensions a and b. In this case, a=3,10mm and b=1,55mm. The TE10 mode start to propagate at 48GHz the next mode, TE01, at 97GHz, then the waveguide is analyzed at 76GHz, frequency in which it will work. Drawing the waveguide in HFSS, it is shown so

Fig. 5 – Rectangular waveguide. HFSS model

The inner rectangular prism is assigned to vacuum, and the side faces are assigned perfect E boundaries. Two wave ports are assigned on the rectangles at -z/2 and +z/2 , using the first propagation mode. The next figure shows the E-field along the waveguide

Fig. 6 – Electric field inside the waveguide

Analyzing the Scattering parameters from 40 to 90GHz, it is got

Fig. 7 – S parameters for the rectangular waveguide

where it can be seen that the first mode starts to propagate inside the waveguide at 48,5GHz.

From 97GHz, TE01 mode could be propagated too, it does not interest us, then the analysis is done at 76GHz.

WAVEGUIDE TRANSITIONS

The most common transitions are from waveguide to coaxial, or from waveguide to microstrip line, to be able to use the propagated energy in another kind of applications. For this, a probe is placed in the direction of the E-field, coupling its energy on the probe. (TE01 mode is in Y-axis)

Fig. 8 – Probe location

The probe is a quarter wavelength resonant antenna at the desired frequency. In X-axis, E-field maximum value happens at x=a/2, while to find the maximum in Z-axis, the guide is finished in a short circuit. So, E-field is null on the guide wall, being maximum at a quarter guide wavelength which is

${\lambda_g}=\dfrac {\lambda}{\sqrt {1-\left( \dfrac {f_c}{f} \right)^2}}$

and in our case, at 76GHz, λ is 3,95mm and λg, 5,11mm. Then, the probe length will be 0,99mm and the shortcircuit distance, 2,56mm.

In coaxial transitions, it is enough to put a coax whose internal conductor protrude λ/4 at λg/4 from the shortcircuit. But in microstrip transitions dielectrics are used as support of the conductor lines, then it should be kept in mindpor the dielectric effect, too.

Our transition can be modeled in HFSS by assigning different materials. The probe is built on Rogers RO3003 substrate, with low dielectric constant and losses, making the transition to microstrip. The lateral faces and the lines are assigned to perfect E boundaries, and form of the substrate, to a RO3003 material. The waveguide inside and the transition cavity is assigned to vacuum. In the extreme face of the transition, a wave port is assigned.

Fig. 10 – Rectangular waveguide to microstrip transition

Now, the simulation is done analyzing the fields and S parameters.

Fig. 11 – E-field on the transition

and it can be seen how the E-field couples to the probe and the signal is propagated along the microstrip.

Fig. 12 – Transition S parameters

Seeing the S parameters, we can see that the least loss coupling happens at 76÷78GHz, our working frequency.

OTHER DEVICES IN WAVEGUIDES: THE MAGIC TEE

Among the usual waveguide devices, one of the most popular is the Magic Tee, a special combiner which can be used like a divider, a combiner and a signal adder/subtractor.

Fig. 13 – Magic Tee

Its behavior is very simple: when an EM field is fed by port 2, the signal is divided and in phase by ports 1 and 3. Port 4 is isolated because its E-plane is perpendicular to the port 2 E-plane. But if the EM field is fed by port 4, it is divided into ports 1 and 3 in phase opposition (180deg) while port 2 is now isolated.

Using the FEM simulation to analyze the Magic Tee, and feeding the power through port 2, it is got the next response

Fig. 14 – E-field inside the Magic Tee feeding by the port 2.

and the power is splitted in ports 1 and 3 while port 4 is isolated. Doing the same operation from port 4, it is got

Fig. 15 – E-field inside the Magic Tee feeding by the port 4.

where now port 2 is isolated.

To see the phases, it is used a vector plot of the E-field

Fig. 16 – Vector E-field inside the Magic Tee feeding by the port 2

where it is seen that the field in ports 1 and 3 has the same direction and therefore they are in phase. Feeding from port 4

Fig. 17 – Vector E-field inside the Magic Tee feeding by the port 2

in which it is seen that the signals in port 1 and 3 has the same level, but in phase opposition (180deg between them).

FEM simulation allows us to analyze the behavior of the EM field from different points of view, only changing the excitations. For example, feeding a signal in phase by port 2 and 4, both signals will be added in phase at port3 and will be nulled at port 1.

Fig. 18 – E-field inside the feeding by ports 2 and 4 in phase.

whereas if inverting the phase in port 2 or port 4, the signals will be added at port 1 and will be nulled at port 3.

Fig. 19 – E-field inside the feeding by ports 2 and 4 in phase opposition

and the result is a signal adder/subtractor.

CONCLUSIONS

The object of this post was the analysis of the electrical behavior of the waveguides using a 3D FEM simulator. The advantage of using these simulators is that they allow to analyze with good precision the EM fields on three-dimensional structures, being the modeling the most important part to rightly define the structure to be studied, since a 3D simulator requires meshing in the structure, and this meshing, as it needs a high number of tetrahedra to achieve good convergence, also tends to need more machine memory and processing capacity.
The structures analyzed, due to their simplicity, have not required long simulation time and relevant processing capacity, but as the models become more complex, the processing capacity increases, it it is needed to achieve a good accuracy.

In subsequent posts, another methods to reduce modeling in complex structures will be analyzed, through the use of planes of symmetry that allow us to divide the structure and reduce meshing considerably..

REFERENCES

1. Daniel G. Swanson, Jr.,Wolfgang J. R. Hoefer; “Microwave Circuit Modeling Using Electromagnetic Field Simulation”; Artech House, 2003, ISBN 1-58053-308-6
2. Paul Wade, “Rectangular Waveguide to Coax Transition Design”, QEX, Nov/Dec 2006

# Using the Three-Dimensional Smith Chart

The Smith Chart is a standard tool in RF design. Developed by Phillip Smith in 1939, it has become the most popular graphic method for representing impedances and solving operations with complex numbers. Traditionally, the Smith Chart has been used as 2-D polar form, centered at an unit radius circle. However, the 2D format has some restrictions when the active impedances (oscillators) or stability circles (amplifiers) are represented, since these ones usually leave the polar chart. Last years, three-dimensional Smith Chart has become popular. Advances in 3D rendering software make it easy to use for design. In this post, I will try to show the handling of the three-dimensional Smith Chart and its application for a low-noise amplifier design.

When Phillip Smith was working at Bell Labs, he have to match one antenna and he was looked for a way to solve the design graphically. By means of the mathematical expressions that define the impedances in the transmission lines, he got to represent the impedance complex plane by circles with constant resistances and reactances. These circles made it easier for him to represent any impedance in a polar space, with the maximum matching placed in the center of the chart and the outer circle representing the pure reactance. Traditionally, Smith’s Chart has been represented in polar form as shown below

Fig. 1 – Traditional Smith’s Chart

The impedance is normalized calculating the ratio between the impedance and the generator impedance. The center of the chart is pure unit resistance (maximum matching) while the peripheral circle that limits the chart is the pure reactance. The left end of the chart represents the pure short circuit and the right end, the pure open circuit. The chart was then very popular to be able to perform calculations for matching networks with transmission lines using a graphical method. However, the design difficulties with the chart happened when active impedances were analyzed, studying amplifiers stability and designing oscillators.

By its design, the chart is limited to the impedances with positive real part, but it could represent, extending the complex plane through the Möbius transformation, impedances with negative real part [1]. This expanded chart, to the negative real part plane, can be seen in the following figure

Fig. 2- Smith’s Chart expanded to active impedances

However,this chart shows two issues: 1) although it allows to represent all the impedances, there is a problem with the complex infinity, so it remains limited and 2) the chart has large dimensions that make it difficult to us in a graphic environment, even in a computer-aided environment. However, the extension is needed when the amplifier stability circles are analyzing, since in most of cases the centers of these circles are located outside the passive impedance chart.

In a graphical computer environment, representing the circles is already performed by the software itself through the calculations, being able to limit the chart to the passive region and drawing only a part of the circle of stability. But with oscillators still have the problem of complex infinity, which could be solved through a representation in a Riemann’s sphere.

RIEMANN’S SPHERE

The Riemann’s sphere is a mathematical solution for representing the complete complex plane, including infinity. The entire complex surface is represented on a spherical surface by a stereographic projection of this plane.

Fig. 3 – Projection of the complex plane on a sphere

In this graphic form the southern hemisphere represents the origin, the northern hemisphere represents infinity and the equator the circle of unitary radius. The distribution of complex values in the sphere can be seen in the following figure

Fig. 4 – Distribution of complex values in the sphere

So, it is possible to represent any complex number on a surface easy to handle.

SMITH’S CHART ON A RIEMANN’S SPHERE

Since Smith’s Chart is a complex representation, it can be projected in the same way to a Riemann’s sphere [2], as shown in the following figure

Fig. 5 – Projection of the Smith’s Chart on a Riemann’s sphere

In this case, the northern hemisphere shows the impedances with positive resistance (passive impedances), in the southern hemisphere, the impedances with negative resistance (active impedances), in the eastern hemisphere, the inductive impedances, and in the western one the capacitive impedances. The main meridian shows the pure resistive impedance.

Thus, when we wish to represent any impedance, either active or passive, it can be represented at any point in the sphere, greatly facilitating its drawing. In the same way, we can represent the stability circles of any amplifier without having to expand the chart. For example, if we want to represent the stability circles for one transistor, which parameters S at 3GHz are the next

S11=0,82/-69,5   S21=5,66/113,8   S12=0,03/48,8  S22=0,72/-37,6

its representation in the conventional Smith’s Chart is

Fig. 6 – Traditional representation for stability circles

while in the three-dimensional chart it is

Fig. 7 – Stability circles on the 3D chart

where both circles can be seen, a fraction in the northern hemisphere and the other one in the south. Thus, its representation has been greatly facilitated.

A PRACTICAL APPLICATION: LOW NOISE AMPLIFIER

Let’s see a practical application of the 3D chart matching the previous amplifier with the maximum stable gain and minimum figure of noise, at 3GHz. Using traditional methods, and knowing the transistor parameters which are the next

S11=0,82/-69,5   S21=5,66/113,8   S12=0,03/48,8  S22=0,72/-37,6

NFmin=0,62  Γopt=0,5/67,5 Rn=0,2

S-parameters are represented in the3D Smith’s chart and the stability circles are drawn. For a better representation 3 frequencies are used, with a 500MHz bandwidth.

Fig. 8 – S-parameters and stability circles for the transistor (S11 S21 S12 S22 Input Stability Circle Output Stability Circle)

It can be seen that S-parameters as well as the stability circles in both the conventional Smith’s chart and 3D one. In the conventional Smith’s chart, the stability circles leave the chart.

One amplifier is unconditionally stable when the stability circles are placed in the active impedance area of the chart, in the southern hemisphere, under two conditions: if the circles are placed in the active region and do not surround the passive one, the unstable impedances are located inside the circle. If the circles surround the passive region, the unstable impedances are located outside the circle.

.

Fig. 9 – Possible cases for stability circles in the active region

In this case, since part of the circles enters on the passive impedances region, the amplifier is conditionally stable.Then the impedances that could unstabilize the amplifier are placed inside the circles. This is something that cannot be seen clearly in the three-dimensional chart yet, the app does not seem to calculate it and would be interesting to include in later versions, because it would greatly facilitate the design.

Let’s match now the input for the minimum noise. For this, it is needed to design a matching network to transform from 50Ω to reflection coefficient Γopt, being its normalized impedance Zopt=0,86+j⋅1,07. In the app, opening the design window and writing this impedance

Fig. 10 – Representation of Γopt

Using now the admittance, we translate in the circle of constant conductance until the real part of the impedance is 1. This is down by estimation and a 0,5 subsceptance is got. It should be increased 0,5 – (- 0,57) = 1.07 and this is a shunt capacitor, 1,14pF.

Fig. 11 – Translating to circle with real part 1.

Now it is only needed to put a component that makes zero the reactance, when the resistance is constant. As the reactance is -1.09, the added value should be 1.09, so that the reactance is zero. This is equivalent to a series inductor, 2,9nH.

Fig. 12 – Source impedance matched to Γopt

Once calculated the input matching network for the lower noise figure, we recalculate the S-parameters. Being an active device, the matching network transforms the S parameters, which are:

S11=0,54/-177   S21=8,3/61,1   S12=0,04/-3,9  S22=0,72/-48,6

and which are represented in the Smith’s chart to get the stability circles.

Fig. 13 – Transistor with matching network to Γopt and stability circles.

The unstable regions are the internal regions, so the amplifier remains stable.

Now the output matching network is got for maximum stable gain, and the ouput reflection coefficient S22=0,72/-48,6 should be loaded by ΓL (S22  conjugate), translating from 50Ω to ΓL=0,72/48,6. This operation is performed in the same way that input matching network. By doing the complete matching , S parameters are recalculated, with input and oputput matching networks. These are

S11=0,83/145   S21=12/-7.5   S12=0,06/-72,5  S22=0,005/162

The gain is 20·log(S21)=21,6dB, and the noise figure, 0,62dB (NFmin). Now it is only represented these parameters in the three-dimensional chart to get the stability circles.

Fig. 14 – Low noise amplifier and stability circles

In this case, the stable region in the input stability circle is inside and in the otuput stabiliy circle is outside. Due to both reflection coefficients, S11 y S22 are into the stable regions, then the amplifier is stable.

CONCLUSIONS

In this entry I had the first contact with the three-dimensional Smith’s chart. The object was to study its potential with respect the traditional chart in microwave engineering. New advantages are observed in this respect in that it is possible to represent the infinite values ​​from the Möbius transform to a Riemann’s sphere and thus having a three-dimensional graphical tool where practically all passive and active impedances and parameters which can be difficult to draw in the traditional chart as stability circles.

In its version 1, the app, which can be found on the website 3D Smith Chart / A New Vision in Microwave Analysis and Design, shows some design options and configurations, although some applications should be undoubtedly added In future versions. In this case, one of the most advantageous applications for the chart, having studied the stability circles of an amplifier, is the location of the stability regions graphically. Although this can be solved by calculation, the visual image is always more advantageous.

The app has a user manual with examples explained in a simple way, so that the designer becomes familiar with it immediately. In my professional opinion, it is an ideal tool for those of us who are used to using Smith’s chart to perform our matching network calculations.

REFERENCES

1. Müller, Andrei; Dascalu, Dan C; Soto, Pablo; Boria, Vicente E.; ” The 3D Smith Chart and Its Practical Applications”; Microwave Journal, vol. 5, no. 7, pp. 64–74, Jul. 2012
2. Andrei A. Muller, P. Soto, D. Dascalu, D. Neculoiu and V. E. Boria, “A 3D Smith Chart based on the Riemann Sphere for Active and Passive Microwave Circuits,” IEEE Microwave and Wireless Components. Letters, vol 21, issue 6, pp 286-288, june, 2011
3. Zelley, Chris; “A spherical representation of the Smith Chart”; IEEE Microwave, vol. 8, pp. 60–66, July 2007
4. Grebennikov, Andrei; Kumar, Narendra; Yarman, Binboga S.; “Broadband RF and Microwave Amplifiers”; Boca Raton: CRC Press, 2016; ISBN 978-1-1388-0020-5